This invention relates to a switching power supply circuit which includes a power factor improvement circuit.
The applicant of the present invention has proposed various power supply circuits which include a resonance type converter on the primary side. Also various power supply circuits wherein a power factor improvement circuit for improving the power factor is provided for a resonance type converter have been proposed by the applicant of the present invention.
FIG. 10 is a circuit diagram showing an example of a switching power supply circuit constructed in accordance with the invention having been applied for patent by the present applicant. The power supply circuit is constructed such that a power factor improvement circuit for improving the power factor is provided for a self-excited current resonance type switching converter.
The power supply circuit shown in FIG. 10 includes a bridge rectification circuit Di for full wave rectifying a commercial ac power supply AC. In this instance, a rectification output obtained by rectification by the bridge rectification circuit Di is charged into a smoothing capacitor Ci through a power factor improvement circuit 20, and a rectified smoothed voltage Ei corresponding to the level equal to the ac input voltage VAC is obtained across the smoothing capacitor Ci.
An inrush current limiting resistor Ri is inserted in the rectification current path of the rectification smoothing circuit (Di, Ci) so that inrush current to flow into the smoothing capacitor, for example, when supply of power is started, may be suppressed.
In the power factor improvement circuit 20 shown in FIG. 10, a filter choke coil LN--high speed recovery type diode D1--choke coil LS connected in series are inserted between a positive output terminal of the bridge rectification circuit Di and a positive terminal of the smoothing capacitor Ci.
A filter capacitor CN is interposed between the anode side of the high speed recovery type diode D1 and the positive terminal of the smoothing capacitor Ci to form a low-pass filter of a normal mode together with the filter choke coil LN.
In the power factor improvement circuit 20, a terminal of a primary side series resonance circuit which is hereinafter described is connected to a connection point between the cathode of the high speed recovery type diode D1 and the choke coil LS so that a switching output obtained by the dc resonance circuit may be fed back.
It is to be noted that a power factor improvement operation of the power factor improvement circuit 20 is hereinafter described.
The power supply circuit includes a self-excited current resonance type converter which uses the rectified smoothed voltage Ei which is a voltage across the smoothing capacitor Ci as operating current.
The current resonance type switching converter includes a pair of switching elements Q1 and Q2 formed from bipolar transistors, connected in a half bridge connection as seen in FIG. 10 and interposed between the positive electrode side connection point of the smoothing capacitor Ci and the ground.
Starting resistors RS1 and RS2 are interposed between the collector and the base of the switching elements Q1 and Q2, respectively. A pair of resistors RB1 and RB2 connected to the base of the switching elements Q1 and Q2 set base current (drive current) of the switching elements Q1 and Q2. A pair of clamp diodes DD1 and DD2 are interposed between the base and the emitter of the switching elements Q1 and Q2, respectively. The clamp diodes DD1 and DD2 form current paths for clamp current which flows between the base and the emitter of the switching elements Q1 and Q2 within periods within which the switching elements Q1 and Q2 are off.
A pair of resonance capacitors CB1 and CB2 form series resonance circuits for self-excited oscillation (self-excited oscillation driving circuits) together with drive windings NB1 and NB2 of a drive transformer PRT (Power Regulating Transformer), which are described subsequently, and determine switching frequencies of the switching elements Q1 and Q2.
The drive transformer PRT is provided to drive the switching elements Q1 and Q2 and variably control the switching frequencies to perform constant voltage control. The drive transformer PRT shown in FIG. 10 is formed as an orthogonal saturable reactor on which the drive windings NB1 and NB2 and a resonance current detection wiring ND are wound and a control winding NC is wound in a direction orthogonal to the windings.
An end of the drive winding NB1 of the drive transformer PRT is connected to the base of the switching element Q1 through a series connection of the resistor RB1 and the resonance capacitor CB1, and the other end of the drive winding NB1 is connected to the emitter of the switching element Q1. An end of the drive winding NB2 is connected to the ground, and the other end of the drive winding NB2 is connected to the base of the switching element Q2 through a series connection of the resistor RB2 and the resonance capacitor CB2. The drive winding NB1 and the drive winding NB2 are wound such that they may generate voltages having the polarities opposite to each other.
An insulation converter transformer PIT (Power Isolation Transformer) transmits switching outputs of the switching elements Q1 and Q2 to the secondary side. An end of the primary winding N1 of the insulation converter transformer PIT is connected to a connection point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 through the resonance current detection wiring ND so that a switching output may be obtained.
The other end of the primary winding N1 is connected to a connection point between the cathode of the high speed recovery type diode D1 in the power factor improvement circuit 20 and the choke coil LS through a series resonance capacitor C1.
In this instance, the series resonance capacitor C1 and the primary winding N1 are connected in series. Thus, a primary side series resonance circuit for making operation of the switching converter operation of the current resonance type is formed from a capacitance of the series resonance capacitor C1 and a leakage inductance component of the insulating converter transformer PIT including the primary winding N1 (series resonance winding).
On the secondary side of the insulating converter transformer PIT shown in FIG. 10, a center tap is provided for the secondary winding N2, and rectification diodes D01, D02, D03 and D04 and smoothing capacitors C01 and C02 are connected in such a manner as seen in FIG. 10. By the connection, two sets of full wave rectification circuits including a set of the [rectification diodes D01 and D02 and smoothing capacitor C01] and another set of the [rectification diodes D03 and D04 and smoothing capacitor C02] are provided. The full-wave rectification circuit formed from the [rectification diodes D01 and D02 and smoothing capacitor C01] produces a dc output voltage E01, and the full-wave rectification circuit formed from the [rectification diodes D03 and D04 and smoothing capacitor C02] produces another dc output voltage E02.
It is to be noted that, in this instance, the dc output voltage E01 and the dc output voltage E02 are branched and inputted also to a control circuit 1. The control circuit 1 utilizes the dc output voltage E01 as a detection voltage and utilizes the dc output voltage E02 as an operation power supply to the control circuit 1.
The control circuit 1 supplies dc current whose level is varied, for example, in response to the level of the dc output voltage E01 on the secondary side as control current to the control winding NC of the drive transformer PRT to perform constant voltage control in such a manner as hereinafter described.
In a switching operation of the power supply circuit having the construction described above, when a commercial ac power supply is made available first, for example, starting current is supplied to the bases of the switching elements Q1 and Q2 through the starting resistors RS1 and RS2, respectively. The switching elements Q1 and Q2 are controlled so that, for example, if the switching element Q1 is switched on first, then the switching element Q2 is controlled so that it is switched off. Then, as an output of the switching element Q1, resonance current flows through the resonance current detection winding ND.fwdarw.primary winding N1.fwdarw.series resonance capacitor C1. The switching elements Q1 and Q2 are controlled so that, around a time at which the resonance current decreases to zero, the switching element Q2 is switched on and the switching element Q1 is switched off. Then, resonance current flows in the reverse direction to that described above through the switching element Q2. Thereafter, a self-excited switching operation wherein the switching elements Q1 and Q2 are alternately switched on is performed.
As the switching elements Q1 and Q2 alternately repeat on-off operations using the terminal voltage of the smoothing capacitor Ci as an operating power supply in this manner, drive current having a waveform proximate to a resonance current waveform is supplied to the primary winding N1 of the insulating converter transformer PIT while an alternating output is obtained at the secondary winding N2.
The constant voltage control by the drive transformer PRT is performed in the following manner.
For example, if the secondary side dc output voltage E01 varies into a rising direction in response to the ac input voltage level, a load variation or the like, then also the level of the control current to flow through the control winding NC is controlled so as to increase in response to the rise of the secondary side dc output voltage E01 as described hereinabove.
While the drive transformer PRT is inclined to approach a saturation condition by an influence of magnetic flux generated in the drive transformer PRT and this acts to drop the inductance of the drive windings NB1 and NB2, the condition of the self-excited oscillation circuit is varied so that the switching frequency may be raised.
While the switching frequency in the power supply circuit is set in a frequency region higher than the resonance frequency of the series resonance circuit of the series resonance capacitor C1 and the primary winding N1 (upper side control), if the switching frequency rises as described above, then the switching frequency is spaced away from the resonance frequency of the series resonance circuit. Consequently, the resonance impedance of the series resonance circuit with respect to the switching output increases.
Since the resonance.impedance increases in this manner and this suppresses the drive current to be supplied to the primary winding N1 of the primary side series resonance circuit, the secondary side output voltage is suppressed, thereby achieving constant voltage control.
It is to be noted that the constant voltage control system by such a method as described above is hereinafter referred to as "switching frequency control system".
The power factor improvement operation by the power factor improvement circuit 20 is such as follows.
In the construction of the power factor improvement circuit 20 shown in FIG. 10, the switching output supplied to the series resonance circuit (N1, C1) is fed back to the rectified current path through an inductive reactance (magnetic coupling) which the choke coil LS itself has.
With the switching output fed back in such a manner as described above, an alternating voltage of the switching period is superposed on the rectified current path. By the superposed component of the alternating voltage of the switching period, an operation of interrupting the rectified current in the switching period is obtained at the high speed recovery type diode D1. By the interruption operation, however, also the apparent inductance of the filter choke coil LN and the choke coil LS increases. Consequently, charging current to the smoothing capacitor Ci flows also within a period within which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci.
As a result, an average waveform of the ac input current approaches the waveform of the ac input voltage to increase the continuity angle of the ac input current, and consequently, improvement of the power factor is achieved.
FIG. 11 is a circuit diagram showing another construction example of a switching power supply circuit which can be constructed based on the invention proposed formerly by the applicant of the present application. Also the present power supply circuit includes a current resonance type converter wherein two switching elements are connected in a half bridge connection. However, the driving system for the power supply circuit is a separate excitation system. Also in this instance, the power supply circuit includes a power factor improvement circuit for achieving power factor improvement.
It is to be noted that like reference characters are applied to like elements to those of FIG. 10 and description thereof is omitted.
The primary side current resonance type converter shown in FIG. 11 includes two switching elements Q11 and Q12 which are, for example, MOS-FETs.
The drain of the switching element Q11 is connected to a line of a rectified smoothed voltage Ei and the source of the switching element Q11 and the drain of the switching element Q12 are connected to each other while the source of the switching element Q12 is connected to the primary side ground thereby to obtain a half bridge connection of the separate excitation type.
The switching elements Q11 and Q12 are driven for switching by an oscillation drive circuit 2 so that on/off operations thereof may be repeated alternately to interrupt the rectified smoothed voltage Ei to obtain a switching output.
In this instance, clamp diodes DD1 and DD2 connected in such directions as indicated in FIG. 11 are provided between the drain and the source of the switching elements Q11 and Q12.
In this instance, an end of a primary winding N1 of an insulation converter transformer PIT is connected to a connection point (switching output point) between the source and the drain of the switching elements Q11 and Q12 so that the switching output may be supplied to the primary winding N1. The other end of the primary winding N1 is connected through a series resonance capacitor C1 to a connection point between a filter choke coil LN of a power factor improvement circuit 21, which is described below, and the anode of a high speed recovery type diode D1.
Also in this instance, a series resonance circuit for making the switching power supply circuit a circuit of the current resonance type is formed from the capacitance of the series resonance capacitor C1 and a leakage inductance component of the insulation converter transformer PIT including the primary winding N1.
A control circuit 1 in this instance outputs, for example, a control signal of a level corresponding to a variation of a dc output voltage E01 to the oscillation drive circuit 2. The oscillation drive circuit 2 varies, based on the control signal supplied thereto from the control circuit 1, the frequencies of the switching driving signals to be supplied from the oscillation drive circuit 2 to the gates of the switching elements Q11 and Q12 to vary the switching frequency.
Also in the power supply circuit shown in FIG. 11, similarly as in the power supply circuit shown in FIG. 10, the switching frequency is set within a region higher than the series resonance frequency. Then, for example, if the dc output voltage E01 rises, then the control circuit 1 controls the oscillation drive circuit 2 so that the switching frequency may be raised in response to the level of the dc output voltage E01. Consequently, constant voltage control is performed in a similar manner as that described with reference to FIG. 10.
A starting circuit 3 is provided to detect a voltage or current obtained at the rectification smoothing line immediately after the power supply is made available to activate the oscillation drive circuit 2. The starting circuit 3 receives, as an operation power supply, a dc voltage of a low level obtained by rectifying a winding wound additionally on the insulation converter transformer PIT.
In the power factor improvement circuit 21 shown in FIG. 11, a filter choke coil LN and a high speed recovery type diode D1 connected in series are interposed between the positive output terminal of the bridge rectification circuit Di and the positive terminal of the smoothing capacitor Ci. Here, the filter capacitor CN is provided in parallel to the series connection circuit of the filter choke coil LN and the high speed recovery type diode D1. Also in such a connection form as just described, the filter capacitor CN forms a low-pass filter of a normal mode together with the filter choke coil LN.
A resonance capacitor C3 is provided in parallel to the high speed recovery type diode D1. Although detailed description is omitted here, for example, the resonance capacitor C3 forms a parallel resonance circuit, for example, together with the filter choke coil LN and so forth, and the resonance frequency of the parallel resonance circuit is set so as to be substantially equal to the resonance frequency of a series resonance circuit which is hereinafter described. Consequently, an action of suppressing a rise of the rectified smoothed voltage Ei when the load decreases is provided.
In the power factor improvement circuit 21, an end portion of a series resonance circuit (N1, C1) is connected to a connection point between the filter choke coil LN and the anode of the high speed recovery type diode D1 as described hereinabove.
In such a connection scheme as described above, a switching output obtained at the primary winding N1 is fed back to the rectified current path through an electrostatic capacitance coupling of the series resonance capacitor C1. In this instance, resonance current obtained at the primary winding N1 is fed back so that it flows to the connection point between the filter choke coil LN and the anode of the high speed recovery type diode D1 so that the switching output may be applied.
Since the switching output is fed back in such a manner as described above, the alternating voltage of the switching period is superposed on the rectified current path, and owing to the superposed alternating voltage of the switching period, an operation of interrupting the rectified current in the switching period is obtained at the high speed recovery type diode D1. Also the apparent inductance of the filter choke coil LN is raised by the interruption operation.
Further, since current of the switching period flows through the resonance capacitor C3, a voltage is generated across the resonance capacitor C3, and the level of the rectified smoothed voltage Ei is lowered by the voltage across the series resonance capacitor C1. Consequently, charging current to the smoothing capacitor Ci flows also within a period within which the rectified output voltage level is lower than the voltage across the smoothing capacitor Ci.
As a result, the average waveform of the ac input current approaches the waveform of the ac input voltage thereby to increase the continuity angle of the ac input current, and also in this instance, power factor improvement is achieved.
In this manner, the power supply circuits shown in FIGS. 10 and 11 can achieve power factor improvement due to the provision of a power factor improvement circuit (20, 21). Since each of the power factor improvement circuits shown in FIGS. 10 and 11 is formed from a small number of parts, they have a merit that power factor improvement can be achieved in a high efficiency, with low noise, with a reduced size and weight and at a low cost.
Here, a relationship between the load power Po and the power factor PF with regard to the power supply circuits shown in FIGS. 10 and 11 is illustrated in FIG. 12. It is to be noted here that a condition when the ac input voltage VAC=100 V is illustrated.
According to FIG. 12, it can be seen that a characteristic that the power factor PF decreases in response to decrease of the load power Po is obtained.
A relationship between the ac input voltage VAC and the power factor PF is illustrated in FIG. 13. Here, characteristics under the conditions of the maximum load power Pomax=120 W and the minimum load power Pomin=40 W are illustrated.
As shown in FIG. 13, it can be seen that the power factor PF decreases in proportion to a rise of the ac input voltage VAC.
The power factor PF under the condition of the minimum load power Pomin=40 W is lower than that under the condition of the maximum load power Pomax=120 W. In short, the characteristic that the power factor PF decreases as the load power decreases as described hereinabove in connection with FIG. 12 is obtained also here.
The characteristics illustrated in FIG. 13 are represented as operation waveform diagrams as seen in FIGS. 14A to 14D.
Here, the ac input voltage VAC and the ac input current IAC under the conditions of the ac input voltage VAC=100 V and the maximum load power Pomax=120 W are illustrated in FIGS. 14A and 14B, and the ac input voltage VAC and the ac input current IAC under the conditions of the ac input voltage VAC=100 V and the minimum load power Pomin=40 W are illustrated in FIGS. 14C and 14D.
Here, if it is assume that the half period of the ac input voltage VAC is 10 ms, then when the load power is the maximum load power Pomax=120 W. the continuity period .tau. of the ac input current IAC actually is approximately 5 ms and the power factor is PF=0.85. On the other hand, when the load power is the minimum load power Pomin=40 W, the continuity period .tau. of the ac input current IAC decreases to approximately 2.5 ms and the power factor drops to approximately PF=0.65. The value of the power factor PF obtained when the load power is the minimum load power Pomin=40 W does not sometimes satisfy a value of the load factor required for actual use.
Since the power factor is dropped by a variation of the ac input voltage or a variation of the load power in this manner, conversely speaking, the ac input voltage or the load condition to the power supply circuits are limited. In short, the power supply circuits have a problem in that apparatus which can adopt the power supply circuits are limited.
More particularly, although the power supply circuits can be adopted, for example, by a color television receiver for which the ac input voltage and the load condition are designated, they cannot be adopted by business apparatus or information apparatus.
Further, it is known that, with the constructions for power factor improvement shown in FIGS. 10 and 11, since they adopt the form that the series resonance circuit on the primary side is connected to the rectified current path of the commercial ac power supply, ripples of the commercial ac power supply period (50 Hz/60 Hz) are superposed on the series resonance circuit. The superposition level of such ripple components increases in proportion to increase of the load power.
It is known that, for example, if it is assumed that the power supply circuits are constructed using required parts selected so that a power factor of approximately PF=0.8 may be maintained under predetermined measurement conditions with which practical use can be provided, then compared with an alternative case wherein no power factor improvement circuit is provided, the ripple voltage level appearing with the secondary side dc output voltage when the load power is maximum increases to approximately 3 to 4 times.
To suppress such increase of ripple components as described above, for example, the power supply circuits shown in FIGS. 10 and 11 actually take such a countermeasure as augmentation of the gain of the control circuit 1 or increase of the capacitance of the smoothing capacitor Ci on the primary side. This, however, gives rise to problems that the cost of part elements increases and that the switching operation is liable to suffer from abnormal oscillation.